High speed source synchronous signaling for interfacing VLSI CMOS circuits to transmission lines

ABSTRACT

A system of the present invention uses small swing differential source synchronous voltage and timing reference (SSVTR and /SSVTR) signals to compare single-ended signals of the same slew rate generated at the same time from the same integrated circuit for high frequency signaling. The SSVTR and /SSVTR signals toggle every time the valid signals are driven by the transmitting integrated circuit. Each signal receiver includes two comparators, one for comparing the signal against SSVTR and the other for comparing the signal against /SSVTR. A present signal binary value determines which comparator is coupled to the receiver output, optionally by using exclusive-OR logic with SSVTR and /SSVTR. The coupled comparator in the receiver detects whether change in signal binary value occurred or not until SSVTR and /SSVTR have changed their binary value. The same comparator is coupled if the signal transitions. The comparator is de-coupled if no transition occurs.

PRIORITY REFERENCES TO PROVISIONAL APPLICATION

This application is a continuation of U.S. patent application Ser. No.10/947,892 filed Sep. 22, 2004, which is a divisional of U.S. patentapplication Ser. No. 09/851,622 filed May 8, 2001, U.S. Pat. No.6,812,767, issued Nov. 2, 2004, which is a continuation of U.S. patentapplication Ser. No. 09/475,087 filed Dec. 30, 1999, U.S. Pat. No.6,255,859 issued Jul. 3, 2001, which is a divisional of U.S. patentapplication Ser. No. 09/057,158 filed Apr. 7, 1998, U.S. Pat. No.6,160,423, issued on Dec. 12, 2000, which claims priority to provisionalapplication Ser. No. 60/078,213 filed Mar. 16, 1998, each of which ishereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to computer signal communication, andmore particularly to an integrated circuit interface and method for highspeed block transfer signaling of data, control and address signalsbetween multiple integrated circuits on a bus or point-to-point withreduced power consumption.

2. Description of the Background Art

Semiconductor integrated circuits used in digital computing and otherdigital applications often use a plurality of Very Large ScaleIntegration (VLSI) interconnected circuits for implementing binarycommunication across single or multi-segmented transmission lines.Conventional transmission lines include traces, which are formed on asuitable substrate, such as a printed circuit board. Each transmissionline may be designed, for example, using so-called micro-strip tracesand strip line traces to form a transmission line having acharacteristic impedance on the order of about 50–70 ohms.Alternatively, each transmission line may have its opposite endsterminated in their characteristic impedance. The output load on adriver for such a transmission line may be as low as 25–35 ohms.

To consume reasonable power, high frequency signaling requires smallamplitude signals. For a receiver to detect voltage swings (e.g., 0.8 vto 1.2 v) easily in a noisy environment like GTL, HSTL, SSTL or RAMBUS,the current must also be very large (e.g., on the order of 50 to 60milliamps per driver). A typical receiver uses a comparator with avoltage reference (VREF) signal configured midway between input highvoltage (VIH) and input low voltage (VIL). The VREF signal is a highimpedance DC voltage reference which tracks loosely with power suppliesover time, but cannot respond to instantaneous noise. Conventionally,High Output Voltage (VOH) and Low Output Voltage (VOL) denote signalsemerging from the transmitting source, and VIL and VIH denote signalsarriving at the input of the receiving device, although they can beconsidered the same signal.

FIG. 1A is a block diagram illustrating a prior art receiver 10 usingRAMBUS technology. The system 10 includes a pad 100 coupled via signallines 103 to internal input receivers 110. A VREF signal 105 is coupledto each internal receiver 110. VREF is generated from the power supply.Usually, the DC value of the power supply varies by five percent (5%).FIG. 1B is a timing diagram 125 illustrating an example signal relativeto a high reference voltage (VREFh) and a low reference voltage (VREFl).The VREFh and VREFl values typically depend on power supply variationused to generate the VREF signal. The large voltage swing, i.e., thedifference between a high voltage signal (VIH) and a low voltage signal(VIL), and stable signal levels above and below the VREF signal arerequired for reliable detection of signal polarity. The voltage swing ofcurrent single-ended signaling technologies is conventionally around 0.8v.

FIG. 1C is a block diagram illustrating schematics of a prior artreceiver 150 using RAMBUS technology. The receiver 150 samples the levelof input signal 167 and of the VREF signal 154 until the signal reachesa stable level, at which time the pass gates 160 and 165 turn off. Oncethe pass gates 160 and 165 turn off, the sense gate 172 is enabled toeliminate current injection. FIG. 1D is a timing diagram 175illustrating operation of the receiver 150 for an example signal. Thereceiver 150 samples the input reference and input signal until thesignal reaches a stable level, e.g., a low logic level (VIL), and, whilethe input signal is stable, the receiver 150 senses the value of theinput signal. As stated above, for reliable signal detection, the signalvoltage swing must be fast to allow all the receivers 150 to sample astable signal with an adequate margin for set-up and hold time. Thisvoltage swing should occur in less than 30% of the minimum cycle time toallow margin for signal skew, set-up and hold-times. As the minimumcycle time reduces below 1 nanosecond, the margins reduce for signalskew, set-up time and hold-time, with the additional burden on thedriver current in a high capacitance loading environment operating athigh frequency. Low voltage differential signaling (LVDS) used by IEEEP1596.3 can overcome these problems by using a 250 mv voltage swing atthe expense of running complimentary signals. Running complementarysignals inevitably increases the pin count and package size.

Accordingly, there is a need for low power drivers and reliablereceivers for high frequency operation of a large number of single-endedsignals in existing technology for low cost VLSI digital systems.

SUMMARY AND OBJECTS OF THE INVENTION

A system of the present invention uses small swing differential sourcesynchronous voltage and timing reference signals (SSVTR and /SSVTR) tocompare single-ended signals of the same swing generated from the sameintegrated circuit for high frequency signaling. It will be appreciatedthat “/” is being used to indicate a logical NOT. All signals areterminated with their characteristic impedance on both ends of thetransmission lines. SSVTR and /SSVTR toggle every time the valid signalsare driven by the transmitting integrated circuit. Each signal receiverincludes two comparators, one for comparing the signal against SSVTR andthe other for comparing the signal against /SSVTR. A present signalbinary value determines which comparator is coupled, optionally by usingexclusive-OR logic with SSVTR and /SSVTR. Until SSVTR and /SSVTR havechanged their binary value, the coupled comparator in the receiverdetects whether a change in signal binary value occurred. Again, it willbe appreciated that SSVTR and /SSVTR change their binary value everytime the signal can change its binary value. SSVTR and /SSVTR arepreferably synchronized with the signal.

The method of the present invention includes the steps of obtaining anoscillating source synchronous voltage and timing reference and itscomplement (SSVTR and /SSVTR), and receiving an incoming single-endedsignal. The method compares the oscillating reference against theincoming signal by a first comparator to generate a first result, andcompares the complement against the incoming signal by a secondcomparator to generate a second result. The method then selects one ofthe first result or the second result as an output signal based on theprevious signal. The step of selecting one of the results includescomparing the output signal to the reference (SSVTR) and to thecomplement (/SSVTR). The step of selecting further includes manipulatingthe output signal from the previous signal towards the first result orsecond result, based on the comparator which is currently coupled. Ifthe incoming signal changes, the step of selecting includes maintainingthe same comparator coupled. If the incoming signal stays the same, thestep of selecting includes de-coupling the currently coupled comparatorand coupling the other comparator. The method then allows the circuit tostabilize.

The system and method advantageously eliminate the need for a highimpedance VREF signal for comparison of small swing single-endedsignals. This reduces the need for three distinct voltage levels (theoutput high level, output low level and the VREF level) to two distinctvoltage levels (the output high level and the output low level).Eliminating VREF reduces necessary voltage swing and accordingly reducespower consumption. Using a receiver with dual comparators allowscoupling of the receiver to the same comparator when the signal changesevery cycle. Only one comparator is coupled based on the current binaryvalue of the signal and SSVTR. The system has an individually adjustabledelay for each receiver to couple or de-couple the comparator, therebyreducing the effect of skew during transmission of source synchronoussignals. The system may have multiple differential source synchronousvoltage and timing reference signals to compare multiple single-endedsignals in the same integrated circuit such as a microprocessor orsystem controller that has many signals. The system and method providedifferential signaling benefits in a single-ended signaling system.

Using the same concept, the system may have bi-directional complementarysource synchronous voltage and timing reference signals to comparebi-directional single-ended signals. The system may have a driver ortransmitter for controlling the signal slew rate to be a substantialportion the total signal period, thereby reducing output current. Thesystem may have internal impedance matching circuitry such as pull-upresistors or grounded gate p-channel for matching the characteristicimpedance of the transmission line on both ends of a point-to-pointconnection between CPU and cache or CPU and system controller. Thesystem has a dual comparator circuit to convert a single-ended bus withtwo complimentary signals to be transmitted and received with comparablenoise immunity of differential bus for internal data bus of memory,processor or other wide data bus type integrated circuits. The systempreferably has variable device size of the transmitter with slowturning-on and slow turning-off to have similar slew rates for allsignals in each group of SSVTR and /SSVTR and plurality of signals whichare transmitted together.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a block diagram illustrating a prior art RAMBUS-basedreceiver.

FIG. 1B is a timing diagram illustrating signal levels of the FIG. 1Aprior art receiver.

FIG. 1C is a schematic diagram illustrating another prior artRAMBUS-based receiver.

FIG. 1D is a timing diagram illustrating operation of the FIG. 1C priorart receiver.

FIG. 2A is a perspective view block diagram illustrating a system with amaster and slave devices in accordance with the present invention.

FIG. 2B is a block diagram illustrating the FIG. 2A system havingtransmission lines with impedance matching resistors at the ends.

FIG. 3A is a timing diagram illustrating the differential referencesignals SSVTR and /SSVTR relative to signal sense times.

FIG. 3B is a timing diagram illustrating SSVTR and /SSVTR relative to asingle-ended signal.

FIG. 4 is a high level schematic illustrating single-ended signalreceivers.

FIG. 5 is a flowchart illustrating a method of communicating signalsfrom a transmitter across a transmission line to a receiver.

FIG. 6A is a schematic diagram illustrating a slow turning-on and slowturning-off driver for all signals.

FIG. 6B is a schematic diagram illustrating drivers having adjustablesignal slew rates and skew between signals.

FIG. 7A is a schematic diagram illustrating a FIG. 4 single-ended signalreceiver in a first embodiment.

FIG. 7B is a schematic diagram illustrating a FIG. 4 single-ended signalreceiver in a second embodiment.

FIG. 7C is a schematic diagram illustrating a FIG. 4 single-ended signalreceiver in a third embodiment.

FIG. 7D is a schematic diagram illustrating a FIG. 4 single-ended signalreceiver in a fourth embodiment.

FIG. 8A is a schematic diagram illustrating circuit details of the SSVTRto /SSVTR comparator of FIG. 4.

FIG. 8B is a schematic diagram illustrating circuit details of the/SSVTR to SSVTR comparator of FIG. 4.

FIG. 9 is a schematic diagram illustrating receivers with individuallyadjustable delays to eliminate skew during transmission.

FIG. 10 illustrates signal waveforms and skew between them.

FIG. 11 is a perspective view of a hard-wire layout of the FIG. 2system.

FIG. 12A is a block diagram illustrating a point-to-point system inaccordance with this invention.

FIG. 12B is a block diagram illustrating the FIG. 12A point-to-pointconnection having impedance-matching grounded gate p-channel devicesinside the integrated circuit.

FIG. 13A is a perspective view block diagram illustrating aunidirectional signaling system and a bi-directional signaling system ona single integrated circuit.

FIG. 13B is a perspective view block diagram illustrating four signalingsystems on a single integrated circuit.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention provides a signaling system and method forhigh-speed communication on multiplexed bus or point-to-pointconnections between multiple VLSI devices with lower power consumptionrelative to current methodology of interfacing single-ended signals. Thesignaling system can be used to connect multiple memory devices with amultiplexed bus to a memory controller for block transfer of data,addresses and control information. By using multiple buses, devices suchas DRAMs, cross-point switches, processors, wide SRAMs and Systemcontrollers can be put together to achieve bandwidths above fourgigabytes/sec. Virtually all of the signals needed for computer or otherdigital systems can be sent over this bus. Persons skilled in the artwill recognize that all devices like CPUs in the computer system needinclude the methodologies and bus structures of this system.

FIG. 2A is a perspective view block diagram illustrating a system 200with a master device (transmitters) 205 coupled via a bus architecture(transmission lines) 215 to multiple slave devices (receivers) 210 inaccordance with the present invention. An example of a master device 205and slave device 210 pair includes a microprocessor and systemcontroller, or a memory controller and memory device (e.g., DRAM). Asillustrated, the master 205 is configured to communicate, for example,twenty (20) signals including single-ended signals S0 to S17, smallswing complementary source synchronous voltage and timing referencesSSVTR and /SSVTR, power lines (not shown) and ground lines (not shown)in parallel via transmission lines 215 to each slave 210. It will beappreciated that “/” is being used to indicate a logical NOT. Thesignals S0–S17 can be data, control or addresses either multiplexed ornon-multiplexed as defined by the protocol. There may be additionalsignals like clock or initialization for other purposes required by theprotocol or synchronization of system.

As shown in FIG. 3A, the SSVTR and /SSVTR signals toggle every timevalid signals are driven by the master 205. It will be appreciated thatslave 210 may include multiple receivers (405, FIG. 4), wherein eachreceiver 405 includes two comparators, one for comparing the signalagainst SSVTR and the other for comparing the signal against /SSVTR. Apresent signal binary value determines which comparator is coupled tothe output terminal 420, optionally by using exclusive-OR logic withSSVTR and /SSVTR. Until SSVTR and /SSVTR have changed their binaryvalue, the enabled comparator in the receiver 405 detects whether changein signal binary value occurred.

For chip-to-chip communication on a bus or point to point, all signalsare transmitted preferably at substantially the same time from the samechip to another chip or plurality of chips connected on the bus andpreferably have substantially the same loading, swing and slew rate(when the signals are transitioning). Also, for intra-chipcommunication, the signals are driven preferably at substantially thesame time from the same area or block to other areas or other blocks inthe same chip and preferably have substantially the same loading, swingand slew rate (when the signals are transitioning)

To facilitate extremely high data transmission rates over this externalbus, the bus cycles are initiated when SSVTR is low (i.e., /SSVTR ishigh). All block transfer begins during the cycle when SSVTR is low andends with SSVTR going low to ease presetting the receiver 405 for thelast binary value of the signal. This allows burst transfers of evennumber of bits. When the signals need to change direction (due to themultiplex nature of signals), one or more dead cycles may be requiredfor settling down the bus due to propagation delays or settling of SSVTRand /SSVTR, when they are bi-directional.

FIG. 2B is a block diagram illustrating the system 200 (FIG. 2A) havingtransmission lines 215 with external impedance matching resistors 220having termination resistance equal to their characteristic impedance,which is preferably between 50–70 ohms, at the ends. The terminationvoltage is labeled VTT, which is preferably around 1.8 v for a 2.5 voperating voltage (for VCC of 2.5V and VSS of 0V). The nominal voltageswing is preferably set less than one volt, preferably less than 40% ofthe supply voltage, and most preferably set at 500 mv. Therefore, asshown in FIG. 3A, the output high voltage (VOH) is 1.8 v and output lowvoltage (VOL) is 1.3 v.

FIG. 3A is a timing diagram illustrating the complementary referencesignals SSVTR and /SSVTR relative to signal sense times. SSVTR initiatesat VOL and /SSVTR initiates at VOH. In the first cycle, the master 205drives all the low going signals including /SSVTR to VOL at the sametime and the termination resistances 220 pull up SSVTR to VOH. Thesingle-ended signals that are high are held at VOH by the terminatingresistances. Proper sense time, i.e., time to sense the logic level ofan input signal, is after the transition junction of SSVTR and /SSVTRand before the stable time, i.e., when the SSVTR or /SSVTR reachessteady state at VIH or at VIL. The SSVTR and /SSVTR preferably haveequal rise and fall times, wherein each rise and fall time isapproximately half of a cycle time of either reference.

FIG. 3B is a timing diagram illustrating SSVTR and /SSVTR relative to asingle-ended signal. The single-ended signal begins equal to /SSVTR at ahigh voltage, and then transitions with /SSVTR to a low voltage. Thesingle-ended signal then remains at a low voltage, thereby becomingequal to SSVTR, and then transitions with SSVTR to a high voltage. Thesingle-ended signal then remains at a high voltage, thereby becomingequal to /SSVTR.

FIG. 4(4-1 and 4-2) is a high level schematic illustrating asingle-ended signal slave 210, having a receiver 405 for each signalline 215. Each signal receiver 405 has two comparators 410, onecomparator 410 a for comparing an incoming single-ended signal “SNx” toSSVTR and the other comparator 410 b for comparing SNx to /SSVTR. Bothof the comparators 410 have output terminals selectively coupled viaswitches 415 to an output terminal 420. It will be appreciated that theoutput signal (SN) to the output terminal 420 is preferably a full railsignal (0V to 2.5V).

As stated above, SSVTR is initially set to VOL and /SSVTR and SNx areinitially set to VOH. SN is initially set to a full rail high outputvoltage. Accordingly, the comparator 410 a amplifies high voltage SNxminus low voltage SSVTR, thereby providing a high output signal. Thecomparator 410 b amplifies high voltage SNx minus high voltage /SSVTR,providing a noise-amplified unknown output signal. Switch 415 selectionis controlled by exclusive-OR (XOR) logic gates 425. More particularly,XOR gate 425 a compares a full rail SSVTR amplified signal (VT) againstoutput signal SN, and generates a control signal for controlling switch415 a. XOR gate 425 b compares full rail /SSVTR (/VT) against outputsignal SN, and generates a control signal for controlling switch 415 b.In this initial state, only SSVTR and accordingly VT are low, therebycausing XOR 425 a to drive switch 415 a closed. Accordingly, thecomparator 410 a output (high) reaches output terminal 420. XOR 425drives switch 415 b open, thereby preventing the entry of the unwantedoutput signal from comparator 410 b. Receiver 405 is stable.

Following the example illustrated in FIG. 3B, the single-ended signalSNx transitions to a low voltage. As always, SSVTR and /SSVTR transitionopposite to one another. Accordingly, as soon as SSVTR and /SSVTRachieve a predetermined difference (preferably 250 mV) therebetween, VTand /VT transition. Similarly, as soon as SSVTR and SNx transition to apredetermined difference (preferably 250 mV) therebetween, the output ofcomparator 410 a also transitions (to a low output voltage). It will beappreciated that the path from external signal SNx to the generation ofoutput signal SN and the path for full rail signal VT and /VT generationpath each include one comparator 410 or 435 and two inverters 430 or440. Thus, each XOR 425 will receive new input signals based on thespeed of the comparison by the comparators 410 and 435. In this example,as evident by the example timing diagram of FIG. 3B, SSVTR and /SSVTRwill achieve a predetermined difference at the same time that SSVTR andSNx achieve the same predetermined difference. Accordingly, the XOR 425a will continue to receive differential inputs, thereby maintaining thesame switch 415 a closed and enabling the low output voltage ofcomparator 410 a to pass to output terminal 420. Receiver 405 is stillstable.

Still following the example of FIG. 3B, the single-ended signal SNx doesnot transition. As always, SSVTR and /SSVTR transition relative to oneanother. Accordingly, currently enabled comparator 410 a continues todrive a low output voltage. When SSVTR and /SSVTR achieve apredetermined difference relative to one another, but before SSVTRreaches the same voltage as SNx (thereby avoiding the possibility of anundetermined state of the output signal), the XOR 425 a switches off andthe XOR 425 b switches on. It will be appreciated that, from the time/SSVTR began to rise, comparator 410 b could drive a low output voltage.Receiver 405 is still stable.

Each receiver 405 can easily detect and amplify very small signals onthe order of 100–250 mV. If the transition has occurred in thesingle-ended signal SNx, the output signal SN has the new level oppositeto its previous signal level. Since both SSVTR (or /SSVTR) andsingle-ended signals have transitioned, the same comparator 410 is stillcoupled to the signal output terminal. If the single-ended signals SNxhave not transitioned, then the signal output SN does not change, thecomparator 410 coupled at the start of the transition is de-coupled fromthe output after the SSVTR and /SSVTR receiver has amplified their newbinary state (VT & /VT), and the other comparator 410 which has opposite/SSVTR (or SSVTR) is coupled to provide the signal output. The oldoutput level is thereby restored.

It will be appreciated that a receiver 405 may be implemented withoutusing XORs. This may be implemented by using the known polarity of SSVTRand /SSVTR in the initial cycle and all single-ended signals startinghigh. The SSVTR and /SSVTR transition in every cycle. Thus, theirpolarity in every cycle may be determined by examining the system clockin a synchronous system and defining cycle start in even clock cycles(i.e., SSVTR is low in the even clock cycle and /SSVTR is high). Then,only the output signal “SN” is monitored to couple and de-couple thecomparators 410 based upon whether output signal SN changes state everycycle or not. If output signal SN changes state, the coupled comparatoris left alone. If the output signal SN does not change, the coupledcomparator is de-coupled and the other comparator is coupled and so on.

It will be further appreciated that a system embodying the inventionenables all signals to be connected to low impedance sources, enablesall signals to present voltage and noise conditions virtually similar todifferential signaling in noise immunity, and enables reduction ofvoltage swing compared to other single-ended signaling technologies likeRAMBUS, HSTL or GTL. The small swing of 0.5 v implemented in thisexemplary embodiment allows for very high signal rates with much lowerpower consumption as compared to other existing single-ended signalingtechnologies. Further, it will be appreciated that each receiver 405amplifies the single-ended signals SNx during the transition of thesignals without the need of a conventional clock or other timing signalexcept SSVTR, /SSVTR and their amplified versions VT and /VT.

FIG. 5(5-1 and 5-2) is a flowchart a method 500 of communicating signalsfrom a master 205 across a transmission line 215 to a receiver 405.Method 500 begins with the master 205 in step 505 setting SSVTR to VOLand all single-ended signals (/SSVTR and SNx) to VOH, and in step 510setting all single-ended receiver outputs (SN) to a full rail high. Thereceiver 405 in step 515 couples the comparator 410 a, which comparesSSVTR against each single-ended signal SNx, to the output terminal 420of the receiver 405. The receiver 405 in step 517 lets all signals onthe transmission lines settle down. Steps 505–517 are referred to assystem initialization.

The master 205 in step 520 simultaneously drives SSVTR and /SSVTR totheir opposite states and all single-ended signals SNx to their desiredlevels. The receiver 405 in step 530 compares the single-ended signalSNx against SSVTR and /SSVTR in respective comparators 410. The receiver405 in step 540 determines whether the single-ended signal transitioned.If so, then the receiver 405 in step 545 passes the result to the outputterminal 420, and keeps the same comparator 410 coupled to the terminal420. If not, then the receiver 405 in step 550 decouples the previouscomparator 410, couples the other comparator 410 to the output terminal420, and keeps the same output signal (SN). The transmitter 405 in step555 determines whether the signal burst continues. If so, then method500 returns to step 520. Otherwise, method 500 ends.

FIG. 6A is a schematic diagram illustrating a slow turning-on and slowturning-off master 205 for a single-ended signal in a first embodimentreferred to as transmitter 600. The transmitter 600 includes an NMOSpull down device 605 coupled to a transmission line 610 for accuratelytailoring the output swing to 500 mv below VTT. The NMOS pull downdevice 605 includes a pull down NMOS transistor T1 having its sourcecoupled to the transmission line 610, its drain coupled to ground, andits gate coupled to skew control circuitry 620. The skew controlcircuitry 620 includes a CMOS inverter, comprising two transistors T2and T3, coupled between two resistors, R1 and R2. The input to the CMOSinverter is coupled to a signal control device 625. For example, togenerate SSVTR or /SSVTR, the signal control device 625 may be anoscillator. It will be appreciated that the amount of pull down can beadjusted using a register (not shown) and a serial pin (not shown)during initialization to set the correct voltage swing for any processor device variations. Other methods like using feedback techniques tocontrol is shown in Hans Schumacher, et al., “CMOS SubnanosecondTrue-ECL output buffer,” J. Solid State Circuits, Vol, 25 (1), pp.150–154 (February 1990) may also be used. Maintaining the current at 20ma and having parallel terminations of 50 ohms on both ends of thetransmission line 610 (as controlled by R1 and R2) generates a 500 mvswing under all conditions. To have slow rise and fall times on theoutput and to minimize reflections, signal coupling and terminationnetwork switching noises, the skew control circuitry 665 controls thepull down transistor TI to turn on and turn off slowly. The preferredslew rate is 1.6 ns/volt with transition times of 0.8 ns for 500 mv.

For a uniformly transitioning ramp-like signal, the preferred slew rateof signals is four times the sum of two inverter delays and anexclusive-OR delay in a given technology. In 0.25μ CMOS technology withan operating voltage of 2.5V, the inverter delay is 50 picoseconds andthe exclusive-OR delay is approximately 120 picoseconds. Thus, thepreferred slew rate is approximately 880 picoseconds. For signalstransmitted above the rate of 600 MHz, the signal slew rate ispreferably less than 110% of the signal rate. The preferred slew ratefor exponential signals is slightly faster if the signal reaches 75% ofits final value earlier than ¾ of the transition time. The differentialsignals preferably cross half way through the voltage transition. Ataround ¾ of the way through the voltage transition, the signals have adifference of about 250 mv which can be converted quickly to a largeswing signal. To avoid noise amplification and to prevent signalcoupling to the receiver output upon receipt non-transitioningsingle-ended signals, the transition time between 75% and the finalsignal value is preferably higher than the sum of two inverter delaysand the exclusive-OR delay. It will be appreciated that the slew ratecan go as fast as it takes amplified noise to reach the output of thecomparator 410 whose output is coupled to the output terminal 420. Thatis, upon receiving a non-transitioning signal, the switches 415 switchstate before the comparator output changes state based on noiseamplification. The output of the currently coupled comparator 410approaches an undetermined (noise amplified) state. The switches 415must switch states before the undetermined output becomes available. Itwill be further appreciated that device mismatches, manufacturingtolerances and signal reflection will affect the speed at which theoutput of the comparator 410 reaches the undetermined state. As thetechnology improves, gate delays, faster slew rates and faster signalrates will be achievable.

FIG. 6B is a schematic diagram illustrating master 205 having adjustablesignal slew rates and skew between signals, in another exemplaryembodiment referred to as transmitter 650. Transmitter 650 includes anNMOS pull down device 655 coupled to the transmission line 610 foraccurately tailoring the output swing to 500 mv below VTT. The NMOS pulldown device 655 includes a pull down NMOS transistors 660 connected inparallel, each having its source coupled to the transmission line 610,its drain coupled to ground, and its gate coupled to skew controlcircuitry 665. The skew control circuitry 665 includes a CMOS inverter,comprising two transistors T2 and T3, coupled between two sets 670 and675 of parallel-connected resistors. The input to the CMOS inverter iscoupled to the signal control device 625. The resistor sets 670 and 675tune the rise and fall times. It will be appreciated that the rise andfall times are preferably as symmetric as possible to have midpointcrossover of all signals and sensing of all signals by the differentialreceivers to occur simultaneously. Achieving symmetry and setting theslew rate and output swing can be achieved during the testing phase byblowing fuses (not shown) or during initialization on board by setting aregister (not shown).

It will be appreciated that the signal transition times may be slightlyhigher than the signal rate. In some heavily loaded buses, the swing canbe increased to take care of transmission losses, still presenting 500mv for the receiver 210 to sense easily. It will be further appreciatedthat various slew rates, exponential transition times and voltage swingsare possible based on technology, loading, and receiver acquisition andresolution delays. Even transition times slightly higher than signalrate are possible with transitioning signals reaching 90 to 95% percentof their final value, while bursting. Also during testing the skewbetween single-ended signals and SSVTR and /SSVTR is adjusted using NMOSpull down size and resistors in the gate prior to it, using well knowntechniques like laser fuse blowing or setting the register code toachieve the signal waveform shape as shown in FIG. 10. As shown in FIG.10, all single-ended signals SNx should be coincident or less than 50psec ahead of the SSVTR and /SSVTR transition. This skew may be adjustedafter testing to be in this range.

FIGS. 7A–7D illustrate alternative embodiments of each signal receiver405 of FIG. 4. It will be appreciated that the comparators 410 ofreceiver 405 need to operate during every cycle, requiring smallacquisition and resolution delays, taking no input current and injectingno current back into signal lines. The common differential amplifiersatisfies all these requirements. Referring to FIG. 7A, the receiver 210uses dual differential amplifiers 702, one differential amplifier 702 afor comparing the signal SNx to SSVTR and the other differentialamplifier 702 b for comparing the signal SNx to /SSVTR. Forcompleteness, a brief review of differential amplifiers 702 is provided.The differential amplifier 702 is always enabled. Based on channelsizes, when the SSVTR voltage is higher than the SNx voltage, morecurrent is driven across the PMOS transistor T10, thereby pulling theoutput voltage at node 707 high (close to VCC or 2.5V). When the SSVTRvoltage is less than the SNx voltage, more current is drawn across theNMOS transistor T11, thereby pulling the output voltage at node 707 low(close to VSS or 0V). The differential amplifier converts 0.5V (smallswing) input to a large swing (0V to 2.5V) output.

The outputs of the differential amplifiers are amplified and inverted byan inverter 704, pass through CMOS transmission gates 706 and are tiedtogether at node 708. The transmission gates 706 are selectivelyoperated depending on the amplified state of previous signal (SN)exclusively-ORed with an amplified state of SSVTR or /SSVTR, i.e. VT or/VT respectively. The exclusive-OR is designed to be stable withoutglitches for, small timing variations between SN, VT and /VT reachingtheir respective logic levels.

Various embodiments are shown. FIG. 7A illustrates an always enableddifferential amplifiers with only the transmission gates beingselectively enabled for small device count and higher speed asalternative embodiment 700. FIG. 7B illustrates a differential amplifierand the transmission gates being enabled or disabled simultaneously asalternative embodiment 720. FIG. 7C illustrates a differentialamplifiers being enabled by the same exclusive-OR for lower power, fastdisabling of transmission gates during transition of exclusive-OR outputand slow enabling of the transmission gates after the exclusive-OR issettled as alternative embodiment 740. FIG. 7D illustrates a P-channeldifferential amplifiers with 1.2V termination voltage for lower powerapplications as alternative embodiment 760. All differential amplifiergates can be disabled for power reduction when the receiver or when thedevice is not selected or the device is in deep power-down mode. Thedifferential amplifier can be disabled by turning transistor T11 off.

By using a 1.2 v termination and receiver 405 as shown in FIG. 7D, thepower consumption can be further reduced by another 33%. That is, thevoltage swing will be from 1.2V to 0.7V, allowing decent margins fromground bounce and lower power consumption for portable systems. Theoperating frequency can be comparable with less number of devices on thebuses, which is common with portable devices for smaller form factor.The transmitter 205 can still be an NMOS pull down T1 or parallelconnection of NMOS pull downs 660. Receiver operation is similar exceptthe differential amplifier 702 becomes a mirror image, therebyincreasing the gate capacitance on signals going into the P-channel gatefor comparable performance by approximately two times due to theincreased device size of the P-channel. Other configurations ofdifferential amplifiers, which convert small swing differential signalsto large swing differential signals quickly, may alternatively be usedinstead of the differential amplifiers shown. One skilled in the artwill recognize that another embodiment can use two different VTTs, onefor signals equal to 1.8 v with 500 mv swing and another for oscillatingreference signals equal to 1.7V with 300 mv swing. All signalstransition at the same time and have similar rise and fall times. Thesame transmitter and receiver pair can manage the multiple VTT system.

It will be appreciated that the DC bias point of each differentialamplifier in the receiver 405 is configured so that the receiver 405output voltage is above half-VCC when both the small swing voltages(single-ended signal SNx and SSVTR or /SSVTR of the enabled differentialamplifier) are close to VIH and below half-VCC when both the small swingvoltages are close to VIL. This DC biasing allows for adequate marginand preservation of output signal SN when the single-ended signal SNxdoes not change state and the SSVTR or /SSVTR of the enableddifferential amplifier is closing the differential signal before it isde-coupled.

Since the receiver 405 operates during the signal transition for a smallswing single-ended signal, the concept of set-up and hold-time from aspecified time after the signal level reaches VIH /VIL or VREF inprevious signaling techniques no longer applies. Also, there is no VREF(reference voltage) for comparison with the signal voltage. Byeliminating the timing necessary for set-up and hold and the timingneeded to enable voltage margins for sensing around VREF, the operatingfrequency is considerably increased with lower power consumption.Further, all receivers 405 are self timed, without the need of a globalclock, allowing the receivers 405 to be adjusted individually forelimination of board or package level transmission skew.

FIGS. 8A and 8B are schematic diagrams illustrating circuit detailscorresponding to comparators 435 of FIG. 4. Each comparator 435 includesa differential amplifier 802 (FIG. 8A) or 852 (FIG. 8B) similar to thedifferential amplifier 702 of FIGS. 7A and multiple inverters 804 (FIG.8A) or 854 (FIG. 8B) in series. The full rail output signals of thecomparators 802 and 852 (VT1, VT2, VT3, /VT1, /VT2 & /VT3) aretransmitted to all the single-ended receivers XORs 425 (FIG. 4).Selection of VT1, VT2 or VT3 is determined based on testing for signalspeed substantially equal to that of the receiver 405 output signal SNgeneration path.

FIG. 9(9-1 and 9-2) is a schematic diagram illustrating receivers 405with individually adjustable delays to eliminate skew duringtransmission and to convert small swing to large swing by comparators410. To tune the operating frequency or voltage swing for optimumperformance, each receiver 405 has a register 905 for storing data toenable delivery of one of the three VT1 & /VT1, VT2 & /VT2 or VT3 & /VT3to the XOR 425 (FIG. 4).

FIG. 11 is a perspective view of a hard-wire layout of a combined master1100 for bi-directional signal communication. The master 1100 includesreceivers 405 and return transmitters 1105 coupled together. Moreparticularly, each single-ended signal received such as signal SO iscoupled to a corresponding receiver 405 such as receiver S0 and to acorresponding transmitter 1105 such as transmitter TO. Preferably, allsingle-ended signals SNx may be grouped together with a single pair ofSSVTR and /SSVTR references. However, persons skilled in the art willrecognize that, for a given operating frequency, SSVTR and /SSVTRloading and signal imbalance reduce the number of signals SNx that canbe grouped together. As shown in FIG. 11, the layout is implemented sothat the capacitances, resistances and inductances on SSVTR, /SSVTR andall single-ended signals SNx are balanced. Also, since SSVTR and /SSVTRgo to all of the receivers 405, the total loading on SSVTR and /SSVTRneeds to be minimized.

By using devices with very low power dissipation and close physicalpacking, the bus can be made as short as possible, which in turn allowsfor short propagation times and high data rates. As shown in FIG. 2B,the resistor-terminated controlled-impedance transmission lines canoperate at signal rates of 1 Ghz (1 ns per cycle). The characteristicsof the transmission lines are strongly affected by the loading caused byintegrated circuits like DRAMs mounted on the bus. These integratedcircuits add lumped capacitance to the lines, which both lowers theimpedance of the lines and decreases the transmission speed. In theloaded environment, the bus impedance is likely to be on the order of 25ohms and the propagation velocity of 7.5 cm/ns. Care should be taken notto drive the bus from two devices at the same time. So for buses lessthan about 12 cm, one dead cycle (e.g., 2 ns) is needed to settle thebus for switching from one driver to another driver. For longer buses,more than one cycle may be needed for the signals to settle down beforea new transmitter can drive the signal. Unlike RAMBUS, the length of thebus does reduce operating frequency in burst mode from the same device.

FIG. 12A is a perspective view block diagram illustrating apoint-to-point system 1200, which includes a bi-directional master 1205coupled via transmission lines 1215 to a bi-directional slave 1210. Thetransmission lines 1215 includes upper signal SNx lines 1220, lowersignal SNx lines 1225 and SSVTR and /SSVTR lines 1230. As illustrated inFIG. 12B is a perspective view block diagram illustrating point-to-pointsystem 1200 incorporating terminating resistances 1235 internally usinggrounded gate P-channel devices. This eliminates the need for space toconnect external resistances and reduces cost. It will be appreciatedthat the terminating resistances 1235 can be implemented using internalresistors instead of grounded gate P-channel devices. Terminating bothends with the appropriate characteristic impedance is preferable forbi-directional signals on a bus. Since intra-chip blocks are physicallyproximate, impedance matching resistances are unnecessary. Small pull-updevices are sufficient. Similarly, when inter-chip connections arephysically proximate, impedance matching resistances can be replacedwith small pull-up devices to reduce cost and power and to maintain thesame slew rate.

It will be appreciated that multiple buses are required for devices likeSLDRAM, DDR SDRAM or DDR SRAMs, where signals are transmitted andreceived simultaneously. FIG. 13A is a perspective view block diagramillustrating a combined unidirectional and bi-directional system 1300for SLDRAM on a single integrated circuit. System 1300 includes a master1305 (e.g., a memory controller) coupled via transmission lines 1315 toslaves 1310 (e.g., SLDRAMs). The master 1305 transmits address andcontrol signals via address and control lines 1320 and 1325,transmits/receives data signals across data lines 1330 and 1335,transmits on SSVTR and /SSVTR lines 1340 a first set of SSVTR and /SSVTRreferences (i.e., SSVTR0 and /SSVTR0) for examining the address andcontrol signals, and transmits a second set of SSVTR and /SSVTRreferences (i.e.,SSVTR1 and /SSVTR1) to the slaves 1310. The address andcontrol portion of the system 1300 manage unidirectional signals neededonly by the slaves 1310. The data portion of the system 1300 isbi-directional based on whether the control signal specified a READ or aWRITE operation.

For an SLDRAM, the 40-bit command and address is sent in a packet offour 10 bit words. SSVTR0 and /SSVTR0, which may be referred to as thesystem differential clock, operates at 500 Mhz. A Phase-Locked Loop (notshown) is used to lock the clock frequency and timing for variousinternal purposes and driving the data output with SSVTR1 and /SSVTR1 onboth edges for a data rate of 1 Ghz. All the high frequency signals areterminated on both ends of the bus with their characteristic impedance.The termination on the memory controller end can include externalresistances, internal resistances or internal grounded gate P-channeldevices, since this memory controller is usually the master and isfixed. Since the number of components (SLDRAMs) 1310 (which operate likeslaves) is variable, components 1310 are preferably terminated byexternal resistors at the end of the transmission lines. The 18 bitbi-directional data bus 1330 and 1335 operates at the same frequency asthe system clock for synchronization and sends data in eight 18-bitwords in four clock cycles (8 ns) or 2.25 gigabytes/sec from a singleSLDRAM. Care is taken to balance the load on SSVTR0 and /SSVTR0 byadding dummy gates and lines to look comparable to SSVTR1 and /SSVTR1.This load balancing makes the slew rate due to loading be similar andallows similar margins for all signals.

When higher bandwidth is required, a system 1350 can use four buses asshown in FIG. 13B. Two separate channels of SLDRAMs 1310 are used with asingle memory controller 1305. This configuration allows 4.5gigabytes/sec peak data bandwidth. Although the system 1350 does notrequire synchronous clocks for the transmitter 1305 or receiver 1310,the system 1350 can use synchronous clocks to transmit data at aparticular time and frequency for ease of testing and usefulness withexisting protocols of synchronous DRAMs and SRAMs. It may be desirableto use an on chip multiplier of a slow clock or an internal ringoscillator to transmit data at high frequency without a high speed clockfor synchronization to reduce noise and system power. It will beappreciated that those skilled in the art can build on the teachings ofthis invention to achieve various size, synchronous or asynchronous,high bandwidth systems.

The foregoing description of the preferred embodiments of the presentinvention is by way of example only, and other variations andmodifications of the above-described embodiments and methods arepossible in light of the foregoing teaching. For example, although thesystem and method have been described as transmitting SSVTR and /SSVTRfrom a master 205 to a receiver 405, one skilled in the art willrecognize that one reference may be sent and the complement generated onthe receiver 405 side. Using the technique with other technologies, suchas bipolar or gallium arsenide, which have similar switching devices andgates, can alternatively be used. Components of this invention may beimplemented using a programmed general purpose digital computer, usingapplication specific integrated circuits, or using a network ofinterconnected conventional components and circuits. The embodimentsdescribed herein are not intended to be exhaustive or limiting. Thepresent invention is limited only by the following claims.

1. In a signal receiver having a first comparator for comparing anincoming signal and a continuously oscillating voltage reference toproduce a first result, a second comparator for comparing the incomingsignal and a complement of the continuously oscillating voltagereference to product a second result, and steering logic for couplingone of the first result and the second result to an output terminal, amethod comprising: defining a time t1 to be a first moment when theincoming signal if transitioning crosses one of the continuouslyoscillating voltage reference and the complement; defining a time t2 tobe a second moment when the incoming signal if not transitioning is atabout a predetermined difference of the one of the continuouslyoscillating voltage reference and the complement; and configuring thesteering logic to cause any change in coupling the one of the firstresult and the second result to the output terminal to occur about at atime between the time t1 and the time t2.
 2. The method of claim 1,wherein the time t1 is earlier than the time t2.
 3. The method of claim1, wherein the predetermined difference is within the range of onlynoise.
 4. The method of claim 1, wherein the predetermined difference asinput to the first comparator is the transition between generating afirst result of logical level one and logical level zero.
 5. The signalreceiver of claim 4, wherein the predetermined difference as input tothe first comparator is the transition between generating a first resultof logical level one and logical level zero.
 6. A signal receiver,comprising: a first comparator for comparing an incoming signal and acontinuously oscillating voltage reference to produce a first result; asecond comparator for comparing the incoming signal and a complement ofthe continuously oscillating voltage reference to produce a secondresult; steering logic for coupling one of the first result and thesecond result to an output terminal, the steering logic for causing anychange in coupling one of the first result and the second result to theoutput terminal, the steering logic for causing any change to occurabout at a time between a time t1 when the incoming signal iftransitioning would cross one of the continuously oscillating voltagereference and it complement and a time t2 when the incoming signal ifnot transitioning would be about at a predetermined difference of theone of the continuously oscillating voltage reference and itscomplement.
 7. The signal receiver of claim 6, wherein the time t1 isearlier than the time t2.
 8. The signal receiver of claim 6, wherein thepredetermined difference is within the range of only noise.